Adjusting the magnitude of a capacitance of a digitally controlled circuit

ABSTRACT

An apparatus comprises a digitally controlled circuit having a variable capacitance and a controller configured to adjust a magnitude of the variable capacitance of the digitally controlled circuit. The digitally controlled circuit comprises a plurality of gain elements, the plurality of gain elements comprising one or more positive voltage-to-frequency gain elements and one or more negative voltage-to-frequency gain elements. The controller is configured to adjust the magnitude of the capacitance by adjusting the gain provided by respective ones of the gain elements in an alternating sequence of the positive voltage-to-frequency gain elements and the negative voltage-to-frequency gain elements.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a Continuation of U.S. patent application Ser. No.14/617,507, filed on Feb. 9, 2015, which claims the benefit of U.S.Provisional Patent Application Ser. No. 62/042,376, filed on Aug. 27,2014. The disclosures of these applications are incorporated byreference herein.

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH

This invention was made with Government support under Contract No.:HR0011-12-C-0087 awarded by the Defense Advanced Research ProjectsAgency (DARPA). The Government has certain rights in this invention.

FIELD OF THE INVENTION

The present application relates generally to oscillators and, moreparticularly to control of an oscillator's frequency.

BACKGROUND

Oscillators are used in various circuits, integrated circuits (chips),and systems. Oscillators often include an inductor, a capacitor, anegative resistance element to maintain oscillation and a variablecapacitor or varactor for controlling the frequency of the oscillator.An oscillator controlled with a digital control word is referred to as adigitally controlled oscillator (DCO). An oscillator controlled with ananalog control voltage is referred to as a voltage controlled oscillator(VCO).

SUMMARY

Embodiments of the invention provide techniques for controlling anoscillator's frequency.

In one embodiment, an apparatus comprises a digitally controlled circuithaving a variable capacitance and a controller configured to adjust amagnitude of the variable capacitance of the digitally controlledcircuit. The digitally controlled circuit comprises a plurality of gainelements, the plurality of gain elements comprising one or more positivevoltage-to-frequency gain elements and one or more negativevoltage-to-frequency gain elements. The controller is configured toadjust the magnitude of the capacitance by adjusting the gain providedby respective ones of the gain elements in an alternating sequence ofthe positive voltage-to-frequency gain elements and the negativevoltage-to-frequency gain elements.

In another embodiment, a phase-locked loop comprises a phase detector, afilter having at least one input coupled to at least one output of thephase detector, a controller having at least one input coupled to atleast one output of the filter, an oscillator having a variablecapacitance, the oscillator having at least one input coupled to atleast one output of the controller, and a divider having at least oneinput coupled to at least one output of the oscillator and at least oneoutput coupled to at least one input of the phase detector. Theoscillator comprises a plurality of gain elements, the plurality of gainelements comprising one or more positive voltage-to-frequency gainelements and one or more negative voltage-to-frequency gain elements.The controller is configured to adjust the magnitude of the capacitanceby adjusting the gain provided by respective ones of the gain elementsin an alternating sequence of the positive voltage-to-frequency gainelements and the negative voltage-to-frequency gain elements.

In another embodiment, a method comprises generating a control voltagefor a digitally controlled circuit having a variable capacitance, thedigitally controlled circuit comprising a plurality of gain elements,providing control signals to the plurality of gain elements by applyingthe control voltage to a first one of the plurality of gain elements andapplying a fixed high voltage or a fixed low voltage to other ones ofthe plurality of gain elements, and adjusting a magnitude of thevariable capacitance of the digitally controlled circuit by adjustingthe control voltage applied to the first gain element and, if the firstgain element is saturated, adjusting the control signals to apply thefixed high voltage or the fixed low voltage to the first gain elementand to apply the control voltage to a second one of the plurality ofgain elements. The first gain element comprises one of a positivevoltage-to-frequency gain element and a negative voltage-to-frequencygain element and the second gain element comprises the other one of apositive voltage-to-frequency gain element and a negativevoltage-to-frequency gain element.

These and other features, objects and advantages of the presentinvention will become apparent from the following detailed descriptionof illustrative embodiments thereof, which is to be read in connectionwith the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a digital phase-locked loop (PLL) architecture, accordingto an embodiment of the invention.

FIG. 2 shows a hybrid PLL architecture, according to an embodiment ofthe invention.

FIG. 3 shows a folding varactor structure, according to an embodiment ofthe invention.

FIG. 4 shows example waveforms for generating a frequency ramp in thefolding varactor structure of FIG. 3, according to an embodiment of theinvention.

FIG. 5 shows varactor voltage-to-frequency transfer curves, according toan embodiment of the invention.

FIG. 6 shows a dual folding varactor structure, according to anembodiment of the invention.

FIG. 7 shows example waveforms for generating a frequency ramp in thedual folding varactor structure of FIG. 6, according to an embodiment ofthe invention.

FIG. 8 shows switching the control of a varactor, according to anembodiment of the invention.

FIG. 9 shows a digital to analog converter (DAC) transfer functionimplementing hysteresis, according to an embodiment of the invention.

FIG. 10 shows a DAC implementation, according to an embodiment of theinvention.

FIG. 11 shows a folding scheme control logic implementation for the dualfolding varactor structure of FIG. 6, according to an embodiment of theinvention.

FIG. 12 shows coarse tuning of a capacitor array, according to anembodiment of the invention.

FIG. 13 shows a high-Q switched capacitor, according to an embodiment ofthe invention.

FIG. 14 shows the effect of a series resistance of a single capacitor onthe Q of a bank of capacitors, according to an embodiment of theinvention.

FIG. 15 shows a folding switched capacitor structure, according to anembodiment of the invention.

FIG. 16 shows example waveforms for generating a frequency ramp in thefolding switched capacitor structure of FIG. 15, according to anembodiment of the invention.

FIG. 17 shows a folding switched capacitor structure with dual switchcapacitor units, according to an embodiment of the invention.

FIG. 18 shows a folding switched capacitor structure with triple switchcapacitor units, according to an embodiment of the invention.

FIG. 19 shows a dual bank folding switched capacitor structure,according to an embodiment of the invention.

FIG. 20 shows the voltage-to-capacitance transfer function for a high-Qswitched capacitor controlled by a single switch, according to anembodiment of the invention.

FIG. 21 shows the voltage-to-capacitance transfer function for a high-Qswitched capacitor controlled with multiple switches having differentthreshold voltages, according to an embodiment of the invention.

FIG. 22 shows a method for controlling a digitally controlled circuit,according to an embodiment of the invention.

FIG. 23 shows a digitally controlled circuit, according to an embodimentof the invention.

DETAILED DESCRIPTION

Illustrative embodiments of the invention will be described herein inthe context of oscillators used in circuits such as PLLs. However, it isto be understood that principles of the invention are not limited solelyto the specific architectures described herein. For example, theinventive techniques can be used in a number of other types of circuitsincluding microprocessors, mm-wave radios, serial links, etc.

Various embodiments provide for controlling the magnitude of acapacitance of a digitally controlled circuit using a “folding”structure. While the folding structure is described below primarily inthe context of oscillators and PLLs, embodiments are not so limited. Byway of example, the folding control structures described herein may beused in a wide variety of other types of digitally controlled circuits,including by way of example filters.

Digital PLLs having low noise and a wide tuning range are desired forvarious applications. The digital control mechanism of an oscillator ina digital PLL is a barrier towards building a digital PLL having lownoise and a wide tuning range.

A high performance, low noise integrated oscillator includes aninductor, a capacitor and a negative resistance element for maintainingoscillation, along with a variable capacitor for controlling thefrequency of the oscillator. DCO architectures may include a DAC used inconjunction with a varactor to form the DCO. DCO architectures may alsouse banks of digitally switched varactors, where each varactor is tiedto a supply. Existing DCO architectures have significant practicallimitations which prevent the implementation of wide tuning range, highfrequency, low noise oscillators.

An ideal DCO structure has a number of desired properties. One desiredproperty is that only a small number of low gain varactors, for exampletwo, should be active at a given time without limiting the total tuningrange of the DCO. A large bank of low gain varactors can be used for arequired tuning range, provided that at any given time most of thevaractors are saturated high or low. Another desired property is thatthe DCO architecture should use low resolution DACs, such as DACs at orbelow an 8-bit level. Higher resolution DACs may be used, but power andarea are compromised when using higher resolution DACs. With regard tothe varactors in the DCO architecture, there are several desiredproperties. For example, there should be no matching requirementsbetween varactors and no simultaneous switching of capacitors that arerequired to match. There should also be no large signal switching ofvaractor control voltages. The DCO structure should also have an overalldigital-to-frequency gain that is reasonably linear, even if thevaractors used in the DCO architecture have low gain regions at the edgeof their respective ranges.

Embodiments of the invention can meet all of the above requirementssimultaneously, overcoming various disadvantages of existing DCOarchitectures which are unable to do so. It is important to note,however, that embodiments are not limited solely to DCO architectureswhich simultaneously meet all of the above requirements.

FIG. 1 shows a digital PLL 100. The digital PLL 100 includes a phasedetector 102, digital control 104, DCO 106, driver 108 and frequencydivider 110. The phase detector 102 detects a phase difference between areference signal, denoted Ref in FIG. 1 and the output of the frequencydivider 110. The frequency divider 110 divides the output frequency ofthe DCO 106 to provide frequency scaling. The output of the phasedetector 102 is input to digital control 104, which controls the DCO 106so as to match the divided output of the DCO 106 with the referencesignal Ref.

FIG. 2 shows a hybrid PLL 200. The hybrid PLL 200 includes phasedetector 202, analog control 203, digital control 204, VCO/DCO 206,driver 208 and frequency divider 210. The phase detector 202 detects aphase difference between the output of the frequency divider 210 and thereference signal Ref. The phase detector 202 in the hybrid PLL 200provides outputs to both analog control 203 and digital control 204.Analog control 203 controls the VCO portion of VCO/DCO 206 while thedigital control 204 controls the DCO portion of VCO/DCO 106. The hybridPLL 200 may be an integer-N hybrid PLL or a fractional-N hybrid PLL.

In some embodiments an apparatus includes an oscillator, such as DCO 106or VCO/DCO 206, having a variable capacitance. The oscillator comprisesa plurality of gain elements, including one or more positivevoltage-to-frequency gain elements and one or more negativevoltage-to-frequency gain elements. A controller is used to adjust themagnitude of the variable capacitance of the oscillator by adjustingcontrol signals for respective ones of the gain elements. The gainelements are controlled in an alternating sequence of the positivevoltage-to-frequency gain elements and the negative voltage-to-frequencygain elements as will be detailed below. In other words, neighboringcontrol elements in the control sequence include one positivevoltage-to-frequency gain element and one negative voltage-to-frequencygain element.

In some embodiments, the apparatus also includes a DAC providing a DACcontrol voltage responsive to control codes supplied by a controllersuch as digital control 104 or digital control 204. A plurality ofswitches are configured to select one of the DAC control voltage, afixed high voltage and a fixed low voltage to apply to correspondingones of the gain elements responsive to control signals provided by thecontroller. In some embodiments described below, the fixed high voltageis a positive supply voltage VDD while the fixed low voltage is a groundvoltage. Embodiments, however, are not limited solely to use with VDDand ground voltages. The controller provides the control signals suchthat one of the gain elements is supplied with the DAC control voltagewhile other ones of the gain elements are supplied with the supply orground voltage.

In other embodiments, the gain elements are arranged into multiple banksof gain elements connected in parallel, and controlled in sequence. Forexample, consider two banks of gain elements where the control signalsfor gain elements in one of the banks are offset from control signalsfor gain elements in the other one of the banks. The control signals inthe banks may be offset by half a range of a single one of the gainelements, as will be described in further detail below. In embodimentswith multiple banks of gain elements, each bank may have its own DAC.Continuing with the two bank example, a first DAC provides a first DACcontrol voltage and a second DAC provides a second DAC control voltage.The controller provides control signals to the first and second bankssuch that one gain element in the first bank is supplied with the firstDAC voltage, one gain element in the second bank is supplied with thesecond DAC voltage, and other ones of the gain elements in the first andsecond banks are supplied with one of the positive supply voltage andthe ground voltage.

In some embodiments, the DAC or DACs have transfer functions whichintroduce hysteresis. For example, the transfer function for a DAC maybe configured such that the top two control codes produce the samemaximum voltage and the bottom two control codes produce the sameminimum voltage.

As will be described in detail below, the gain elements may comprisevaractors or high-Q switched capacitors, where Q denotes the qualityfactor. In some embodiments, the oscillator has a bank of two or moredigitally switched high-Q capacitors for coarse band tuning of thecapacitance of the oscillator, where the gain elements are varactorsproviding continuous tuning of the capacitance of the oscillator withinone of the coarse bands.

The partitioning of a frequency range of a VCO between coarse frequencybands and continuous or analog frequency tuning involves a tradeoff. Asmall analog tuning range results in a lower voltage-to-frequency gain,less sensitivity to noise, etc. but has a limited frequency range overwhich a PLL can maintain a lock. A large analog tuning range allows thePLL to maintain lock over a larger frequency range, but also results inhigher voltage-to-frequency gain. The additional sensitivity due to thehigher voltage-to-frequency gain creates substantial implementationchallenges. Embodiments described below provide a tuning scheme thatbreaks the link between the glitch-free lock range and the requirementfor low gain varactors.

FIG. 3 shows a folding varactor structure 300. The folding varactorstructure 300 may be implemented as part of the DCO 106 in digital PLL100 or the VCO/DCO 206 in hybrid PLL 200.

The folding varactor structure 300 has a bank of 16 low gain analogvaractors, denoted A₀ through A₁₅ in FIG. 3. The low gain varactors A₀through A₁₅ are used in conjunction with a DAC (not shown in FIG. 3) tocontrol the frequency of an oscillator without any matching requirementsbetween the analog varactors and without requiring more than one lowgain varactor to be active at any time. The “even” numbered varactorsA₀, A₂, etc. have positive voltage-to-frequency gain, while the “odd”numbered varactors A₁, A₃, etc. have negative voltage-to-frequency gain.In other embodiments the even numbered varactors may have negativevoltage-to-frequency gain while the odd numbered varactors may havepositive voltage-to-frequency gain.

Although FIG. 3 shows the even and odd numbered varactors beingphysically placed next to one another, this is not a requirement. Theeven numbered varactors and the odd numbered varactors need not bephysically placed next to one another, as long as the sequence in whichpositive voltage-to-frequency varactors and negativevoltage-to-frequency varactors are alternately controlled as describedherein.

A set of switches, shown as multiplexers in FIG. 3 and control logic(not shown in FIG. 3) connect each varactor to one of the DAC voltage,the positive supply voltage (VDD) or the ground voltage (VSS). Thedigital control 104 in digital PLL 100 and the digital control 204 inhybrid PLL 200 may implement the control logic for the folding varactorstructure 300.

A DAC is used to generate a voltage which is applied to one of thevaractors A₀ through A₁₅. All other ones of the varactors are tied toVDD or VSS, corresponding to their respective saturated, low gain state.As shown in FIG. 3, the voltage from the DAC may be provided by a sigmadelta (ΣΔ) modulator. The functionality of the DAC and ΣΔ modulator,along with the RC circuit shown after the DAC/ΣΔ modulator signal in thebottom of FIG. 3, will be described in further detail below with respectto FIG. 9.

Consider a DAC voltage currently supplied to a positivevoltage-to-frequency gain varactor. If the oscillator frequency isrequired to increase, the DAC voltage increases until the voltagereaches VDD. Once the DAC voltage corresponding to full scale isdetected, the current varactor is disconnected from the DAC andconnected to VDD. The DAC is then switched so as to control the nextvaractor in the bank, which has a negative voltage-to-frequency gain. Ifthe oscillator frequency is required to increase further, the DACvoltage ramps down until it eventually reaches VSS, at which point thenegative voltage-to-frequency gain varactor is switched from the DACcontrol voltage to VSS, and the DAC control voltage is applied to thenext varactor in the bank, which is a positive voltage-to-frequency gainvaractor. The DAC voltage then ramps up again to increase the oscillatorfrequency. This process continues through the bank of varactors. If theoscillator frequency is required to decrease, the process is reversed.

FIG. 4 shows waveforms for generating a frequency ramp in the foldingvaractor structure 300. To increase the frequency, the voltages from theDAC and ΣΔ modulator ramp up and down between VDD and VSS. Each positivevoltage-to-frequency gain varactor is initially supplied VSS, while eachnegative voltage-to-frequency gain varactor is initially supplied VDD.As shown in the FIG. 4 waveforms, control voltage va₀ controllingvaractor A₀ starts at VSS, control voltage va₁ controlling varactor A₁starts at VDD, control voltage va₂ controlling varactor A₂ starts atVSS, etc. The frequency ramp starts with the DAC control voltage beingsupplied to varactor A₀. The DAC voltage, and thus control va₀ ramps upto VDD. Once the DAC voltage reaches VDD, control va₀ is switched to VDDand the DAC voltage is supplied to varactor A₁. The DAC voltage, andthus the control va₁ then ramps down to VSS. On reaching VSS, varactorA₁ is switched to VSS and the DAC voltage is supplied to varactor A₂.The DAC voltage, and thus the control va₂ then ramps back up to VDD atwhich point the varactor A₂ is switched to VDD, and the DAC voltage issupplied to varactor A₃. This process continues through varactor A₁₅ oruntil the desired frequency is reached.

It is important to note that although FIG. 3 shows a bank of 16 low gainanalog varactors A₀ through A₁₅, embodiments are not limited solely tohaving 16 varactors. Embodiments may use more or fewer varactors, ormore generally more or fewer gain elements. By way of example, as willbe described in further detail below with respect to FIGS. 6 and 7, someembodiments may use multiple banks of analog varactors.

FIG. 5 shows voltage-to-frequency transfer curves. FIG. 5 shows afrequency-voltage plot and a gain-voltage plot, each showing an idealtransfer function and a typical transfer function. As can be seen, thegain is a non-linear function of the voltage in a typical transferfunction. A consequence of this is that the small signal gain can behighly variable. The largest small signal gain typically occurs in themiddle of a varactor's tuning range. Changes in the center frequency ofan oscillator due to manufacturing variation means that it is difficultto predict which point on the transfer curve will correspond to a givenfrequency. When a DCO is placed in a PLL system, this gain variabilityleads to variability in the PLL's characteristics such as the smallsignal transfer function, phase margin, etc.

To account for the above-described gain variability, some embodimentsutilize multiple banks of gain elements that are simultaneously activebut offset from one another. FIG. 6, for example, shows a dual foldingvaractor structure 600. The dual folding varactor structure shows twobanks of varactors, varactor bank A 601 and varactor bank B 603. Thevaractor bank A 601 has 16 analog varactors denoted A₀ through A₁₅, andthe varactor bank B 603 has 16 analog varactors denoted B₀ through B₁₅.It is to be appreciated that embodiments are not limited solely tovaractor banks containing 16 varactors or more generally gain elements.In addition, embodiments are not limited solely to two varactor banks.The dual varactor folding structure 600, which will be described infurther detail below, may be extended to include more than two banks.This can be advantageous in applications where maintaining a linearvoltage-to-frequency transfer function is critical.

The dual varactor folding structure 600 includes two DACs. DAC A 602provides a control word to varactor bank A 601, while DAC B 604 providesa control word to varactor bank B 603. The DACs 602 and 604 arecontrolled by varactor folding controller 605. The varactor foldingcontroller 605 is an example of the digital control 104 of digital PLL100 and the digital control 204 of hybrid PLL 200. Similar to thefolding varactor structure 300, “even” numbered varactors in each bank,e.g., A₀, A₂, etc. and B₀, B₂, etc., have positive voltage-to-frequencygain while the “odd” numbered varactors in each bank, e.g., A₁, A₃, etc.and B₁, B₃, etc. have negative voltage-to-frequency gain. Again similarto the varactor structure 300 described above, in other embodiments theeven numbered varactors may have negative voltage-to-frequency gainwhile the odd numbered varactors may have positive voltage-to-frequencygain.

As noted above with respect to FIG. 3, although the even and oddnumbered varactors in FIG. 6 are physically placed next to one another,this is not a requirement. The even numbered varactors and the oddnumbered varactors need not be physically placed next to one another, aslong as the sequence in which positive voltage-to-frequency gainvaractors and negative voltage-to-frequency gain varactors arealternately controlled as described herein.

A set of switches, shown in FIG. 6 as multiplexers, are used to connecteach varactor in varactor bank A 601 to one of the DAC voltage providedby DAC A 602, VDD or VSS in response to control signals received fromvaractor folding controller 605. The DAC voltage provided by DAC A 602is controlled based on the 8-bit control (A) received from varactorfolding controller 605. Similarly, a set of switches shown asmultiplexers in FIG. 6 are used to connect each varactor in varactorbank B 603 to one of the DAC voltage provided by DAC B 604, VDD or VSSin response to control signals received from varactor folding controller605. The DAC voltage provided by DAC B 604 is controlled based on the8-bit control (B) received from varactor folding controller 605. At anygiven time, one varactor from varactor bank A 601 is connected to theDAC voltage provided by DAC A 602 and one varactor from varactor bank B603 is connected to the DAC voltage provided by DAC B 604, while theother varactors in varactor bank A 601 and varactor bank B 603 areconnected to VDD or VSS. Thus, at any given time there are two activevaractors—one from varactor bank A 601 and one from varactor bank B 603.

The control code for varactor bank B 603 is offset from the control codefor varactor bank A 601. In some embodiments, the control code forvaractor bank B 603 is offset from the control code for varactor bank A601 by approximately half of the range of a single one of the varactors.More generally, the offset is based on the number of banks of gainelements. For three banks of gain elements, the control codes may beoffset by approximately one third of the range of a single one of thevaractors.

Due to the control code offset, when the voltage applied to the activevaractor in varactor bank A 601 is at the top or bottom of its range,the voltage applied to the active varactor in varactor bank B 603 willbe in the middle of its range. The total code-to-frequency gain is thesum of the gain from varactor bank A 601 and varactor bank B 603. Whenthe gain of varactor bank A 601 is very low due to the active varactorvoltage being close to saturated at supply or ground, the gain ofvaractor bank B 603 will be high as the control voltage for the activevaractor of varactor bank B 603 will be in the center of its range, andvice versa. Consequently, the gain variation versus frequencyillustrated in FIG. 5 is significantly reduced.

FIG. 7 shows waveforms for generating a frequency ramp in the dualfolding varactor structure 600. To increase the frequency, the voltagesfrom DAC A 602 and DAC B 604 ramp up and down between VDD and VSS. Asshown in FIG. 7, the voltages from DAC A 602 and DAC B 604 and/orcorresponding ΣΔ modulators, are offset from one another in the mannerdescribed above. Each positive voltage-to-frequency gain varactor isinitially supplied VSS, while each negative voltage-to-frequency gainvaractor is initially supplied VDD. As shown in the FIG. 7 waveforms,control voltage va₀ controlling varactor A₀ starts at VSS, controlvoltage va₁ controlling varactor A₁ starts at VDD, control voltage va₂controlling varactor A₂ starts at VSS, etc. Similarly, control voltagevb₀ controlling varactor B₀ starts at VSS, control voltage vb₁controlling varactor B₁ starts at VDD, control voltage vb₂ controllingvaractor B₂ starts at VSS, etc. The frequency ramp starts with the DAC A602 control voltage being supplied to varactor A₀. The DAC B 604 controlvoltage also starts as being supplied to varactor B₀, although the DAC B604 control voltage is offset from the DAC A 602 control voltage asdiscussed above.

The DAC A 602 and DAC B 604 control voltages, and thus the correspondingcontrol voltages va₀ and vb₀ ramp up to VDD. Once the DAC A 602 voltagereaches VDD, control va₀ is switched to VDD and the DAC A 602 voltage issupplied to varactor A₁. Due to the above-described offset, the DAC B604 voltage is still controlling vb₀ when the DAC A 602 voltage switchesso as to control varactor A₁ via control va₁. The DAC A 602 voltage, nowcontrolling va₁ for varactor A₁, begins to ramp down to VSS while theDAC B 604 voltage continues to ramp up to VDD until vb₀ reaches VDD.Once the DAC B 604 voltage reaches VDD, the DAC B 604 voltage switchesso as to control varactor B₁ via control vb₁. At this point, the DAC A602 voltage still controls varactor A₁ via control va₁. The DAC A 602voltage and DAC B 604 voltage are then both ramping down towards VSS.

Once the DAC A 602 voltage reaches VSS, the control va₁ is tied to VSSand the DAC A voltage switches so as to control varactor A₂ via controlva₂. While DAC A 602 transitions from control varactor A₁ to controllingvaractor A₂, the DAC B 604 voltage is still controlling varactor B₁ asit ramps down towards VSS due to the offset of the DAC A 602 and DAC B604 voltages. The DAC B 604 voltage continues to ramp down towards VSScontrolling varactor B₁ via control vb₁ while the DAC A 602 voltagebegins to ramp up towards VDD controlling varactor A₂ via control va₂.Once the DAC B 604 voltage reaches VSS, the DAC B 604 voltage switchesso as to control varactor B₂ via control vb₂ and the varactor B₁ is tiedto VSS. At this point, both the DAC A 602 voltage and the DAC B 604voltage are ramping up towards VDD as shown in the waveforms of FIG. 7.This process continues as the frequency increases and the DAC A 602 andDAC B 604 voltages continue to sequentially control varactors in thevaractor bank A 601 and the varactor bank B 603.

Returning to FIG. 6, the dual folding varactor structure 600 may furtherinclude a differentially controlled analog varactor set 606. Thedifferentially controlled analog varactor set 606 can be used to changethe VCO frequency in proportion to the control voltage denoted V_(prop)in FIG. 6. The differentially controlled analog varactor set 606provides an additional way of changing the frequency scheme, which maybe used in addition to the folding varactor control described above. Thedifferentially controlled analog varactor set 606 may be controlled bythe analog control 203 in hybrid PLL 200 as part of the analogproportional path control. In some embodiments, the dual foldingvaractor structure 600 does not include the differentially controlledanalog varactor set 606.

The dual folding varactor structure 600 also includes a set of coarsecontrol fixed capacitors 607. The VCO of the dual folding varactorstructure 600 may include the set of coarse control fixed capacitors 607for coarse band tuning of a magnitude of the variable capacitance of theVCO. Each of the coarse control fixed capacitors 607 may be a digitallyswitched high-Q capacitor unit. Each capacitor unit includes a pair ofcapacitors as shown in FIG. 6. One of the coarse tuning bands may beselected by controlling which of the coarse control fixed capacitors 607is enabled, with the varactor bank A 601 and the varactor bank B 603providing continuous tuning of the magnitude of the variable capacitanceof the VCO within the coarse band selected by the enabled one of thecoarse control fixed capacitors 607. In other embodiments, however,coarse band tuning may be eliminated.

Element 608 in the folding varactor structure 600 is a set of resistorsand switches, which are used to reduce and limit the current in the VCO.

When the DAC voltage switches from one varactor to the next, there is apossibility of a small glitch if the maximum or minimum voltage that theDAC produces is not identical to the voltage applied to the varactorwhen switched to VDD or VSS, respectively. When a DCO is used in a PLL,if the PLL is trying to achieve lock at a frequency corresponding to aboundary between two varactors, this mismatch could result in the PLLswitching back and forth between varactors. This switching can causesmall periodic glitches in the output frequency of the DCO, whichdegrades phase noise performance. FIG. 8 illustrates switching for apositive voltage-to-frequency gain varactor from being controlled by theDAC voltage and being tied to VDD. The top plot in FIG. 8 shows theideal case where the maximum voltage produced by the DAC is identical toVDD. The bottom plot in FIG. 8 shows the voltage error that results whenthe maximum voltage produced by the DAC is not identical to VDD.

To account for the possible voltage error shown in the bottom plot ofFIG. 8, some embodiments introduce hysteresis when switching betweenanalog varactors. FIG. 9 shows a control word applied to a ΣΔ modulator901.

The ΣΔ modulator 901 is not required, but can be used to increase theeffective resolution of DAC 902. For example, consider an 8-bit DACwhich has 256 levels. If the DAC has a reference voltage of 1 volt, thenthe DAC's step size would be 1V/255 or approximately 3.92 mV. If a stepsize smaller than the DAC step size is required, delta-sigma modulator901 can be used to quickly switch the DAC between two adjacent levels sothat the average output voltage is somewhere in between the two voltagelevels that are being switched between. This effectively increases theresolution provided there is a downstream low pass filter that removesthe noise associated with this switching between adjacent DAC levels.The RC filter shown in FIG. 9 after the DAC 902, for example, filtersout this noise. The RC filters shown in FIGS. 3 and 6 provide similarfunctionality.

The ΣΔ modulator 901 may switch the DAC 902 between adjacent levelsusing a variety of patterns. For example, consider a situation in whichthe step size is between levels 6 and 7 of the DAC 902. The delta-sigmamodulator 901 may switch the DAC 902 using a repeating sequence 6, 7, 7,6, 7, 7, etc. to produce an average output for the DAC 902 at a 6.666level. The delta-sigma modulator 901 may alternatively switch the DAC902 using a repeating sequence of 6, 7, 6, 7, etc. to produce an averageoutput for the DAC 902 at a 6.5 level or the delta-sigma modulator 901may switch the DAC 902 using a repeating sequence of 6, 6, 7, 6, 6, 7,etc. to produce an average output for the DAC 902 at a 6.333 level. Aswill be appreciated, various other sequences may be used to obtaindifferent average outputs for the DAC 902.

The plot of FIG. 9 shows the DAC output voltage for control codes N-4,N-3, N-2, N-1, and N. Without hysteresis, the DAC output voltage isdifferent for each control code. With hysteresis, the DAC transferfunction is adjusted such that the top two control codes, N-1 and N,produce the same maximum DAC output voltage. Although not shown, the DACtransfer function may be altered such that the bottom two control codesproduce the same minimum DAC output voltage. This adds hysteresis, suchthat no limit cycles occur due to the glitches described above withrespect to FIG. 8. In some embodiments, hysteresis may be introduced tothe DAC transfer function by having more than the top two and bottom twocontrol codes produce the same maximum and minimum DAC output voltage,respectively. As described above, the DAC transfer function from controlword to analog voltage is not necessarily limited by the quantizationstep size of the DAC 902. The ΣΔ modulator 901 may be used to switchback and forth between the two closest DAC levels so that the correctaverage voltage is generated.

With hysteresis, when the DAC 902 reaches the top or bottom of its rangeand is switched from one varactor to the next, an offset is added to theDAC control such that the DAC starts in the next band at the second fromtop or second from bottom code instead of the top or bottom. The codesecond from the top still produces the maximum DAC voltage, and the codesecond from the bottom similarly still produces the minimum DAC voltage.The extra code switch does not increase the frequency, but increases theamount that the control would have to reduce by before switching back tothe original band. If there is a positive frequency error induced byimperfect switching, then the DAC 902 will be driven back towards thefirst code and not immediately switch back to the original varactor.

In the context of the dual varactor folding structure 600, when a firstone of the varactor bank A 601 and the varactor bank B 603 is switchingbetween varactors, the second bank is by design in the middle of itsrange at or near its highest gain region. When the control code movesback to correct for the switching frequency error, the second bank alsochanges. As the second bank varactor voltage is in the middle of itsrange, it will have significantly more gain than the first bank andhence the frequency error can be corrected for with a relatively smallinput control word change. Provided that correcting the frequency errorcaused by the band switch requires a change of control codecorresponding to less than 1 least significant bit of the DACs 602 and604, the frequency error will not cause the first bank to switch back tothe original varactor.

FIG. 10 shows a DAC implementation 1000 which may be used in thevaractor structure 300 and the folding varactor structure 600. FIG. 10specifically shows an 8-bit DAC implementation. A control word is inputto a binary to Gray code converter 1001. A Gray code is used forencoding such that adjacent control words have a single binary digitthat differs by 1. Flip-flop 1002 references the output of binary toGray code converter 1002 to the system clock. Gray code decode module1003 controls the switches SW1 through SW256 to provide the controlvoltages corresponding to the 256 levels of the 8-bit DAC.

FIG. 11 shows a folding scheme control logic 1100, which may be used inconjunction with the dual folding varactor structure 600. FIG. 11 showsa ΣΔ modulator 1101. In some embodiments, the functionality of the ΣΔmodulator 901 may instead be performed by ΣΔ modulator 1101. In theseembodiments, the outputs of flip flops 1120 and 1122 go directly to DAC602 and DAC 604 in the dual folding varactor structure 600, rather thanto a ΣΔ modulator such as ΣΔ modulator 901.

The input for the folding scheme control logic 1100 is a 12-bit bus. The12 bit input is provided to summer 1105 directly, and to summer 1107 viasummer 1102. Summer 1102 provides the offset between varactor banks. Theoutput of summers 1105 and 1107 are each split into 4 most significantbits (MSBs) and 8 least significant bits (LSBs). The 4 MSBs are used toselect which varactor in each of varactor bank A 601 and varactor bank B603 is controlled by the DAC 602 voltage and the DAC 604 voltage,respectively. The 4 MSBs are provided to binary to gray code converters1115 and 1117, and then to flip-flops 1119 and 1122 before beingprovided to the varactor decoders for varactor bank A 601 and varactorbank B 603. The flip-flops 1119 and 1122 are referenced to a feedback(FB) clock.

As described above, the varactors in each of varactor bank A 601 andvaractor bank B 603 have alternating polarity gains such that the firstvaractor has positive voltage-to-frequency gain, the second varactor hasnegative voltage-to-frequency gain, the third varactor has positivevoltage-to-frequency gain, etc. In varactor bank A 601 and varactor bankB 603, the positive gain varactors are “even” numbered (0, 2, 4, 6 etc.)while the negative gain varactors are “odd” numbered (1, 3, 5, 7, etc.).

When the DAC voltage is being used to control an even numbered varactor,the output of multiplexer 1111 and multiplexer 1114 is set to the 8 LSBsof the input. When the DAC voltage is being used to control an oddnumbered varactor, the output of multiplexer 1111 and multiplexer 1114is modified so that when the input code is minimum, the output code ismaximum and vice versa. Thus, the output of the multiplexers 1111 and1114 when controlling an odd numbered varactor is the full scale minusthe 8 LSBs. Expressed in binary, the output of multiplexers 1111 and1114 when controlling an odd numbered varactor is (11111111)—8 LSBs. Thenotation 8′hff in FIG. 11 is Verilog for 8 bits of hexadecimal “ff”, orin binary 11111111.

In the folding scheme control logic 1100, even detector 1109 and theeven detector 1112 determine whether the 4 MSBs are controlling an oddor even numbered varactor. It is important to note that due to theoffset provided by summer 1102, the 4 MSBs input to even detector 1109and the 4 MSBs input to even detector 1112 need not be the same. Thus,one of the even detectors 1109 and 1112 can determine that an evennumbered varactor in one bank is to be controlled by the DAC voltagewhile the other one of the even detectors 1109 and 1112 determines thatan odd numbered varactor is to be controlled by the DAC voltage in theother bank.

For varactor bank A 601, the 8 LSBs, and the difference between fullscale and the 8 LSBs as determined by summer 1110 are input tomultiplexer 1111. Even detector 1109 provides the control signal formultiplexer 1111 to control which input to the multiplexer 1111 isoutput to binary to gray code converter 1116. The output of binary togray code converter 1116 is input to flip-flip 1120 and to DAC 602. Theflip-flip 1120 is referenced by the FB clock. Similar logic is providedby even detector 1112, summer 1113 and multiplexer 1114 for varactorbank B 603. The elements 1109, 1110, 1111, 1112, 1113 and 1114 form afolding control module 1108 as shown in FIG. 11.

The outputs of flip-flips 1119 and 1121 are fed back to gray code tobinary converters 1104 and 1106, respectively. The output of gray codeto binary converter 1104 is input to summer 1105 along with the 12 bitinput. The summer 1105 provides hysteresis for the DAC A 602 transferfunction in the manner described above with respect to FIGS. 8 and 9.Similarly, the summer 1107 provides hysteresis for the DAC B 604transfer function. Thus, the elements 1103, 1104, 1105, 1106 and 1107form a hysteresis module 1103 as shown in FIG. 11. The folding schemecontrol logic 1100 may be simplified for use with the folding varactorstructure 300. For example, folding varactor structure 300 may becontrolled using only elements 1101, 1105, 1106, 1109, 1110, 1111, 1115,1116, 1119 and 1120 in folding scheme control logic 1100.

The varactor folding structure 300 and the dual folding varactorstructure 600 described above provide a number of advantages relative toconventional techniques. For example, these structures are compatiblewith linearization, are inherently highly glitch-resistant due to themanner in which analog and digital control handoff is managed, and allowfor a wide continuous tuning range coverage without the need for a highgain varactor or high resolution DAC.

In some embodiments, the folding concept described in conjunction withvaractor folding structure 300 and dual varactor folding structure 600may be extended to folding control of high-Q switched capacitors withina main PLL loop. Folding control of high-Q switched capacitors enablesthe elimination of coarse band controls and associated non-idealinfrastructure for wide tuning range, high performance VCOs.

The design of high performance VCOs typically involves a tradeoffbetween the VCO's phase noise performance and the VCO's continuous,glitch free tuning range. High performance VCO designs may beimplemented with a coarse banding approach, which splits the VCO tuningrange between a continuous but low Q part and a discrete high Q part.Continuously tunable varactors, also referred to as analog varactors,generally have lower quality factors than digitally switched capacitors.

VCOs capable of achieving good phase noise performance and a wide tuningrange can be implemented with two sets of varactors, a continuouslytuned analog varactor and one or more digitally switched capacitors inthe form of high-Q capacitors in series with digital switches. However,as temperature and voltage change, the frequency range associated witheach discrete band may move, requiring careful initial band choice andlarge band overlap or loss of lock may occur. Choosing the size of theanalog varactors in such an arrangement involves a tradeoff betweenphase noise of the VCO and the frequency range over which the PLL canmaintain a continuous lock. If the analog varactor size is smallrelative to the total capacitance of the VCO, the frequency range overwhich the PLL can remain locked is small. Small analog varactor size,however, can improve the VCO's phase noise performance because the bulkof the VCO's capacitance can be comprised of fixed, or digitallyswitched high-Q capacitors.

FIG. 12 illustrates an example of the tradeoff that occurs for a coarsetuning capacitor array. FIG. 12 illustrate a 4-bit fine switchedcapacitor array having 8 coarse tuning bands. Continuous tuning isrestricted to within each coarse band, and the bands have significantoverlap with one another. In some embodiments, a folding scheme is usedsuch that a VCO's entire tuning range is accessed within the PLL'sprimary control loop without a phase noise penalty thus implementing ahigh-Q wide tuning and low noise PLL eliminating coarse band control.

FIG. 13 shows an example of a high-Q switched capacitor unit 1300. Asshown in FIG. 13, the high-Q switched capacitor unit 1300 includes apair of capacitors in high-Q switched capacitor unit 1300. At highfrequency, such as frequencies over 10 GHz, phase noise performance islimited by the Q of the capacitors. Fixed metal-to-metal capacitors witha series switch give better phase noise as compared with an analogvaractor. When the enable signal EN is set to 1, the capacitors areconnected to one another. Ideally, when EN=1 the switch, shown in FIG.13 as an NMOS transistor, should have zero on resistance R_(on) and whenEN=0 the switch should have zero off parasitic capacitance C_(off). Inpractice, the switch has some non-zero on resistance R_(on) and somenon-zero parasitic capacitance C_(off). The resistance of the switch maybe reduced by increasing the size of the switch, but this comes at thecost of greater off-state parasitic capacitance. One metric used forswitches in high frequency applications is the R_(on) to C_(off)product. FIG. 13 illustrates the difference between the ideal and actualscenarios.

A digitally controlled switch, such as the NMOS transistor of high-Qswitched capacitor unit 1300, can be gradually transitioned between itson and off state to provide continuous tuning. However, when the controlsignal for the digitally controlled switch is between its extremes theseries loss of the switch will degrade the Q of the switched capacitorunit 1300, resulting in a degradation of phase noise. Folding schemesallow for seamless switching between bands, so the capacitance can bebroken down into a large number of small capacitor units. As long as thesize of the capacitor unit that is transitioning between its on and offstates is a small fraction of the total capacitance, then the entirestructure will have high Q even if one of the capacitors is in a low Qstate. FIG. 14 illustrates this principle, showing the effect of seriesresistance of a single capacitor unit on the Q of a bank of N capacitorunits. FIG. 14 shows a plot of the Q factor for a bank of N capacitorunits over the on-resistance of the switches for capacitor units in thebank, R_(switch). As shown in FIG. 14, there is significant Qdegradation for a small bank of N=10 capacitor units but there isnegligible Q degradation for a large bank of N=256 capacitor units, asthe Q is limited by the on resistance of the capacitor switches R_(min).

It is important to note that embodiments are not limited solely to usewith a bank of 256 digitally switched capacitor units connected inseries. While FIG. 14 shows that such an arrangement generally providesfor negligible Q degradation, different numbers of digitally switchedcapacitors may be used in other embodiments based on an acceptable Qdegradation for a particular application.

FIG. 15 shows a folding switched capacitor structure 1500. FIG. 15 showscapacitor units each comprising a pair of capacitors controlled byswitches sw₀, sw₁, . . . sw₁₂₇. Each switch sw₀, sw₁, . . . , sw₁₂₇ iscontrolled by a corresponding control signal va₀, va₁, . . . va₁₂₇. Theoverall capacitance of the folding switched capacitor structure 1500 issplit into small capacitor units, controlled by alternating NMOS andPMOS switches. As shown in FIG. 15, the “even” digitally switchedcapacitor units 0, 2, 4, etc. are controlled by PMOS switches and areassociated with positive voltage-to-frequency gain while the “odd”digitally switched capacitor units 1, 3, 5, etc. are controlled by NMOSswitches and are associated with negative voltage-to-frequency gain.

Although FIG. 15 shows the even and odd digitally switched capacitorunits physically placed next to one another, this is not a requirement.The even and odd digitally switched capacitor units need not bephysically placed next to one another, as long as the sequence in whichpositive voltage-to-frequency gain ones of the digitally switchedcapacitor units and negative voltage-to-frequency gain ones of thedigitally switched capacitor units are alternately controlled asdescribed herein.

A single thermometer coded capacitor bank is used to cover the entiretuning range, eliminating coarse bands. A folding structure is used togradually transition one of the switches sw₀ through sw₁₂₇ between itson and off state while remaining switches and corresponding capacitorunits are saturated in their on (high Q) or off (high impedance) state.

FIG. 16 shows waveforms for generating a frequency ramp in foldingswitched capacitor structure 1500. The DAC control voltage from ΣΔ rampsup and down between VDD and VSS. Initially, the control voltages appliedto the even PMOS switches are saturated at VSS while the controlvoltages applied to the odd NMOS switches are saturated at VDD. Tobegin, the control va₀ for switched capacitor unit 0 is supplied withthe DAC control voltage. To increase the frequency, the control va₀gradually ramps up to VDD until the capacitor unit 0 is saturated in itshigh impedance state. If the frequency needs to increase further, thecontrol va₀ for capacitor unit 0 is tied to VDD and the control va₁ forcapacitor unit 1 is supplied with the DAC control voltage. The DACcontrol voltage ramps down from VDD to VSS to increase the frequency. Onreaching VSS, the control va₁ is tied to VSS and the control va₂ forcapacitor unit 2 is supplied with the DAC control voltage, which beginsto ramp back up towards VDD. This process continues through capacitorunits 2 through 127, or until the desired frequency is reached.

In the switched capacitor units shown in folding switched capacitorstructure 1500, the resistances shown between the capacitor units have adegrading effect on the Q (de-Qing effect) of the capacitor units. Toaccount for this degrading effect, some embodiments use alternativestructures for the switched capacitor units. FIGS. 17 and 18 show twosuch alternative structures.

FIG. 17 shows folding switched capacitor structure 1700 where eachcapacitor unit has dual switches. Capacitor unit 0 includes two PMOSswitches sw₀ each controlled by control va₀, capacitor unit 1 includestwo NMOS switches sw₁ each controlled by control va₁, etc. The dualswitch arrangement shields the capacitor banks in each capacitor unitfrom the de-Qing effect of the resistance R between the capacitor units.When the switch of the capacitor unit is off, ideally the impedance seenat the switch side of the capacitors would be as large as possible. If aresistor is used to set the DC bias of the nodes connected to thecapacitors, then the switch will reduce the effective off-stateimpedance of the switch/capacitor combination. Nevertheless, it isimportant to provide a DC bias to the drain and source of the switch,particularly when the switched capacitor unit is on. By using twotransistor switches in series as shown in FIG. 17, the transistors canbe provided with a DC bias voltage without the bias resistor reducingthe impedance seen at the capacitor node. When the two transistors ofthe switch are off, the impedance of the DC biasing resistor does notreduce the impedance seen at the capacitor terminal because of the highoff impedance of the switch. The dual switch arrangement shown in FIG.17, however, has additional parasitic capacitance for a given onresistance, as the dual or parallel switch in folding switched capacitorstructure 1700 is twice the size of the single switch used in foldingswitched capacitor structure 1500.

FIG. 18 shows folding switched capacitor structure 1800 where eachcapacitor unit has three switches. Capacitor unit 0 includes three PMOSswitches sw₀ each controlled by control va₀, capacitor unit 1 includesthree NMOS switches sw₁ each controlled by control va₁, etc. The FETswitches create a high resistance in the off state to shield thecapacitor tanks in each unit from the de-Qing effect of the resistance Rbetween capacitor units. The same effect could be achieved without theFET switches by making the resistors bigger. However, in a CMOSmanufacturing process the physical area required to implement a highimpedance resistor is often much larger than the physical area of a FETswitch. As the folding switched capacitor structure 1800 has a largenumber of switched capacitor units, it may be important for the switchedcapacitor units to be implemented in such a way that it is efficient interms of area usage.

FIG. 18 shows each capacitor unit having three FET switches. Forexample, the switched capacitor units each have a “primary” switch (thetop switch) and two “biasing” switches below the primary switch. Whenthe primary switch is on, the biasing switches are also on and theimpedance of the DC biasing network is primarily set by the resistors.The de-Qing effect of the biasing impedance is more important when theprimary switch is off as the off-state impedance should be high. Thus,when the primary switch is off the biasing switches are also off. Theoff-state leakage current from drain to source of the biasing switchesis still sufficient to bias the voltage at the capacitor nodes. Thus,the series switches act to increase the impedance of the DC biasingscheme when the switch is off. The triple switch arrangement shown inFIG. 18, however, has some dependence on device leakage for properoperation.

FIG. 19 shows a dual bank folding switched capacitor structure 1900implemented in a VCO. The dual bank folding switched capacitor structure1900 includes bank A 1901, bank B 1902, differentially controlled analogvaractors 1903 and VCO core 1904. The differentially controlled analogvaractors 1903 operate in a manner similar to the differentiallycontrolled analog varactor set 606 described above with respect to dualfolding varactor structure 600.

Each of bank A 1901 and bank B 1902 includes a series of digitallyswitched capacitor units. FIG. 19 shows each of bank A 1901 and bank B1902 utilizing the dual switch capacitor unit structure described abovewith respect to FIG. 17. Embodiments, however, are not limited solely touse with the dual switch capacitor units. Instead, the bank A 1901 andbank B 1902 may each use the single switch or triple switch arrangementsdescribed above with respect to FIGS. 15 and 18, respectively.

The capacitor units in bank A 1901 and bank B 1902 have multiplexersproviding the control signals to each capacitor unit in the respectivebanks A controller such as digital control 104 in digital PLL 100 ordigital control 204 in hybrid PLL 200 may be used to control themultiplexers for each bank. For bank A 1901 the DAC control voltage fromΣΔ_(a), is provided to one of the capacitor units, while other capacitorunits in bank A 1901 are tied to VDD or VSS. Similarly, for bank B 1902the DAC control voltage from ΣΔ_(b) is provided to one of the capacitorunits, while other capacitor units in bank B 1902 are tied to VDD orVSS. Thus, at any given time one of the capacitor units in bank A 1901is connected to the DAC control voltage from ΣΔ_(a), one of thecapacitor units in bank B 1902 is connected to the DAC control voltagefrom ΣΔ_(b), and the remainder of the capacitor units in bank A 1901 andbank B 1902 are saturated at their high Q or high impedance state bybeing tied to one of VDD and VSS.

The control for each of the banks in dual bank folding switchedcapacitor structure 1900 is similar to the control scheme describedabove with respect to FIG. 16. Each bank's control may be offset fromone another in the manner described above with respect to the dualvaractor bank structure described in conjunction with FIGS. 6 and 7.

Hysteresis may be added into the DAC transfer functions for the switchedcapacitor structures shown in FIGS. 15, 17, 18 and 19 in a mannersimilar to that described above with respect to FIGS. 8 and 9.

Similar to FIG. 15, there is no requirement that the even and odddigitally switch capacitor units in FIGS. 17-19 be physically placednext to one another as long as the sequence in which positivevoltage-to-frequency gain ones of the digitally switched capacitor unitsand negative voltage-to-frequency gain ones of the digitally switchedcapacitor units are alternately controlled as described herein.

FIG. 20 illustrates the voltage-to-capacitance transfer function for ahigh-Q switched capacitor. FIG. 20 shows an NMOS switched capacitorunit. The NMOS switch sw₀ has its gate coupled to control va₀. A MOSswitch, such as NMOS switch sw₀, can turn on too quickly. Thevoltage-to-capacitance transfer function shown in the plots of FIG. 20show that the NMOS switch under appropriate large signal excitationturns from on to off in approximately ¼ of the control voltage change,which may be 250 mV for a 1V process. A PMOS switch has this same issue.Thus, the voltage-to-capacitance gain is highly concentrated in the ¼control voltage change and the voltage-to-capacitance transfer functionis nonlinear. Even in the dual bank folding switched capacitor structure1900, the voltage-to-capacitance structure may be nonlinear. Onesolution is to use four varactor banks, each of which are offset formone another by ¼ of the range of a capacitor unit. This solution,however, requires significantly more hardware.

FIG. 21 illustrates the voltage-to-capacitor transfer function for ahigh-Q switched capacitor unit 2100 having two pairs of capacitors. Thefirst pair of capacitors has an NMOS switch sw_(vt1) and the second pairof capacitors has an NMOS switch sw_(vt2). Control va₀ is supplied tothe gates of switches sw_(vt1) and sw_(vt2). The NMOS switch sw_(vt1)and NMOS switch sw_(vt2) have different threshold voltages. The voltagedifference between the transistor turn on time is equal to thedifference in threshold voltages between NMOS switch sw_(vt1) and NMOSswitch sw_(vt2). The difference in thresholds may be selected to beVDD/4 (250 mV for a 1V process) resulting in a voltage-to-capacitancegain of the aggregate structure that is more linear over the 250 mVrange. Although shown in FIG. 21 with NMOS switches, a high-Q switchedcapacitor unit may also be formed with two PMOS switches havingdifferent threshold voltages.

FIG. 22 shows a method 2200 for controlling a digitally controlledcircuit. The method 2200 beings with generating 2202 a control voltagefor a digitally controlled circuit having a variable capacitance. FIG.23 shows an example of a folding controller 2302 and digitallycontrolled circuit 2304. The folding controller 2302 may perform themethod 2200. The digitally controlled circuit 2304 may be an oscillator,a PLL, a filter, etc. The control voltage may be generated using a DACin the manner described above. The digitally controlled circuit has aplurality of gain elements, including one or more positivevoltage-to-frequency gain elements and one or more negativevoltage-to-frequency gain elements. FIGS. 3, 6, 15, 17, 18 and 19described above show examples of such a digitally controlled circuit.

The method 2200 continues with providing 2204 control signals to theplurality of gain elements of the digitally controlled circuit byapplying the control voltage to a first one of the plurality of gainelements and applying a fixed high voltage or a fixed low voltage toother ones of the plurality of gain elements. As described above withrespect to FIGS. 6 and 19, in some embodiments gain elements of adigitally controlled circuit may be arranged in multiple banks. In theseembodiments, step 2204 involves providing offset control voltages to onegain element in each of the multiple banks.

In step 2206, the magnitude of the variable capacitance of the digitallycontrolled circuit is adjusted by adjusting the control voltage appliedto the first gain element. If the first gain element is saturated andfurther adjustment of the magnitude of the variable capacitance isdesired, the method continues with adjusting 2208 the control signals toapply the control voltage to a second one of the plurality of gainelements while the fixed high voltage or the fixed low voltage isapplied to the first gain element. The first gain element and the secondgain element have opposite polarities, i.e., the first gain element isone of a positive voltage-to-frequency gain element and a negativevoltage-to-frequency gain element while the second gain element is theother one of a positive voltage-to-frequency gain element and a negativevoltage-to-frequency gain element. Steps 2206 and 2208 may be repeatedas necessary to further adjust the magnitude of the capacitance of thedigitally controlled circuit by providing the control voltage to thegain elements in an alternating sequence of positive and negativevoltage-to-frequency gain elements.

Various structures described above may be implemented in integratedcircuits. It is to be appreciated that, in an illustrative integratedcircuit implementation, one or more integrated circuit dies aretypically formed in a pattern on a surface of a wafer. Each such die mayinclude a device comprising circuitry as described herein, and mayinclude other structures or circuits. The dies are cut or diced from thewafer, then packaged as integrated circuits. One ordinarily skilled inthe art would know how to dice wafers and package dies to producepackaged integrated circuits. Integrated circuits, manufactured as aboveand/or in other ways, are considered part of this invention. It is to beunderstood that circuits in some embodiments can be formed acrossmultiple integrated circuits.

It will be appreciated and should be understood that the exemplaryembodiments of the invention described above can be implemented in anumber of different fashions. Given the teachings of the inventionprovided herein, one of ordinary skill in the related art will be ableto contemplate other implementations of the invention. Indeed, althoughillustrative embodiments of the present invention have been describedherein with reference to the accompanying drawings, it is to beunderstood that the invention is not limited to those preciseembodiments, and that various other changes and modifications may bemade by one skilled in the art without departing from the scope orspirit of the invention.

What is claimed is:
 1. A method comprising: generating a control voltagefor a digitally controlled circuit having a variable capacitance, thedigitally controlled circuit comprising a plurality of gain elements;providing control signals to the plurality of gain elements by applyingthe control voltage to a first one of the plurality of gain elements andapplying a fixed high voltage or a fixed low voltage to other ones ofthe plurality of gain elements; and adjusting a magnitude of thevariable capacitance of the digitally controlled circuit by adjustingthe control voltage applied to the first gain element and, if the firstgain element is saturated, adjusting the control signals to apply thefixed high voltage or the fixed low voltage to the first gain elementand to apply the control voltage to a second one of the plurality ofgain elements; wherein the first gain element comprises one of apositive voltage-to-frequency gain element and a negativevoltage-to-frequency gain element and the second gain element comprisesthe other one of a positive voltage-to-frequency gain element and anegative voltage-to-frequency gain element; wherein the plurality ofgain elements comprises two or more banks of gain elements; whereinproviding control signals to the plurality of gain elements comprisesapplying the control voltage to one gain element in each of the two ormore banks and applying the fixed high voltage or the fixed low voltageto other ones of the gain elements in the two or more banks; wherein thecontrol signals for gain elements in the two or more banks are offsetfrom one another; and wherein generating the control voltage for thedigitally controlled circuit comprises: using first and seconddigital-to-analog converters to generate first and seconddigital-to-analog control voltages in response to control codes;selecting one of the first digital-to-analog control voltage, the fixedhigh voltage and the fixed low voltage to connect to corresponding onesof a first one of the two or more banks of gain elements based on thecontrol signals; selecting one of the second digital-to-analog controlvoltage, the fixed high voltage and the fixed low voltage to connect tocorresponding ones of a second one of the two or more banks of gainelements based on the control signals; and adjusting the control signalssuch that one of the gain elements in the first bank is supplied withthe first digital-to-analog control voltage, one of the gain elements inthe second bank is supplied with the second digital-to-analog controlvoltage, and remaining ones of the gain elements in the first bank andthe second bank are supplied with the fixed high voltage or the fixedlow voltage.
 2. The method of claim 1, wherein at least one of the firstdigital-to-analog converter and the second digital-to-analog convertercomprises a transfer function providing hysteresis for the controlcodes.
 3. The method of claim 1, wherein the plurality of gain elementscomprise varactors.
 4. The method of claim 1, wherein the plurality ofgain elements comprise high-Q switched capacitors.
 5. The method ofclaim 1, wherein a magnitude of the offset of the control signals isbased at least in part on the number of banks of gain elements.
 6. Themethod of claim 1, wherein the digitally controlled circuit comprisesone of an oscillator and a filter.
 7. The method of claim 1, wherein thedigitally controlled circuit comprises two or more digitally switchedhigh-Q capacitors for coarse band tuning of the magnitude of thevariable capacitance, the plurality of gain elements comprising aplurality of digitally controlled varactors providing for continuoustuning of the magnitude of the variable capacitance within a givencoarse band.
 8. The method of claim 4, wherein each of the high-Qswitched capacitors comprises a first switched capacitor controlled by afirst transistor and a second switched capacitor controlled by a secondtransistor, and further comprising providing a same control voltage toeach of the first transistor and the second transistor, the firsttransistor and the second transistor having different thresholdvoltages.